Hybrid battery charger

ABSTRACT

A hybrid battery charger is disclosed that includes a linear charger circuit for providing vehicle starting current and battery charging and a high frequency battery charging circuit that provides battery charging current. The linear charger circuit and the high frequency battery charging circuits are selectively enabled to provide vehicle starting current, maximum charging current and optimum efficiency.

RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.15/153,164, filed on May 12, 2016, which is a continuation of U.S. Pat.No. 9,368,269, filed on Oct. 24, 2012, each by Xiao Ping Chen et al. andentitled “Hybrid Battery Charger.” U.S. patent application Ser. No.15/153,164 and U.S. Pat. No. 9,368,269 are hereby incorporated byreference in their entirety.

TECHNICAL FIELD

The present invention relates to a battery charger and more particularlyto a hybrid battery charger that includes a linear charger circuit forproviding vehicle starting current and battery charging duringpredetermined conditions and a high frequency battery charging circuitfor providing battery charging current during different predeterminedconditions; the linear charger circuit and the high frequency batterycharging circuits being selectively enabled to provide vehicle startingcurrent, maximum charging current, and optimum efficiency.

BACKGROUND

Various types of battery charger circuits are known in the art. Forexample, the two most common types of battery charger circuits arelinear and high frequency (also known as switched mode) battery chargercircuits. Both types of battery charger circuits are known to haveadvantages and disadvantages.

Linear charger circuits normally include a transformer, a rectifier anda current regulating element. The primary of the transformer is normallyconnected to an external 120 volt AC power supply. The transformer stepsdown the voltage from the 120 volt AC power supply to an appropriatevoltage for charging a battery, for example 12 volts AC. A rectifier,such as, a full wave rectifier, converts the stepped down AC voltage onthe secondary winding of the transformer to a DC charging voltage. Insome known linear charger circuits, a passive linear element, such as aresistor, is connected in series with the secondary winding of thetransformer to limit the charging current provided to the battery.Linear charger circuits may also include a voltage regulator between thepassive element and the secondary winding of the transformer tostabilize the output voltage. The charging current of such linearcharger circuits is a linear function of the voltage of the 120 volt ACpower supply source.

High frequency charger circuits are also known. An exemplary highfrequency transformer is described in detail in U.S. Pat. No. 6,822,425,hereby incorporated by reference. In general, such high frequencycharger circuits normally are connected to an external 120 volt AC powersupply. The 120 volts AC from the 120 volt AC power supply is rectified,for example, by a full wave rectifier, to generate a DC voltage. The DCvoltage is switched on and off by electronic switching circuitry tocreate a high frequency pulse train, for example, at frequencies from 10KHz to 1 MHz, and applied to a high frequency transformer. The highfrequency transformer steps down the voltage to an appropriate chargingvoltage. This charging voltage is rectified and filtered to provide thedesired DC charging voltage for the battery to be charged.

Regulations governing battery charger efficiencies have been promulgatedby various governmental agencies. For example, the California EnergyCommission has revised their Appliance Efficiency Regulations to includebattery charger circuits. These regulations are set forth in Title 20,Sections 1601-1608 of the California Code of Regulations(“Regulations”). The US Department of Energy has also promulgatedstandards regarding the efficiency of battery chargers in Title 10, Part430 of the Code of Federal Regulations.

Unfortunately, many known conventional linear battery charger andconventional high frequency battery chargers are not known to meet thebattery charger efficiency benchmarks set forth in the standardsmentioned above. Specifically, known linear charger circuits are knownto have efficiency in the range from 50% to 75% at full load, which isbelow the benchmarks set forth in the standards mentioned above. Most ofthe losses are known to be from the transformer.

In order to address this problem, one known linear charger circuit isknown to incorporate a toroidal transformer which has significantlylower losses than bobbin wound transformers. However, there are severaldrawbacks with respect to the use of toroidal transformers. For example,such toroidal transformers require specialized winding equipment aremore labor intensive and have efficiency in the range from 65% to 80% atfull load. In addition, as is the case with most known bobbin woundtransformers, the efficiency of such toroidal wound transformers islower at less than 60% of full load. In fact, at 20% of full load, theefficiency of such toroidal wound transformers can be less than 40%.

High frequency charger circuits can be designed to be 80% to90%+efficient at full load. However, the efficiency of such highfrequency charger circuits is known to be relatively less efficient atless than full load. In addition, high frequency battery chargers areless reliable because of the number of components and the amount ofcurrent through those components in an engine start mode.

As mentioned above, the efficiencies of the linear and high frequencybattery chargers vary as a function of their loading. The regulationsset forth above relate to overall efficiencies. This means that thebattery charger must meet the efficiency benchmarks during allconditions in which the battery charger is connected to a 120 volt ACpower supply. For example, the California regulations specify that theefficiency benchmark must be maintained over a 24 hour period during thefollowing modes of operation:

-   -   A mode when the battery charger is charging a battery.    -   A mode in which the battery charger is providing a trickle        charge to the battery.    -   A mode in which the battery is disconnected from the battery        with the battery charger still connected to the 120 volt AC        power supply.

Moreover, even though the conventional linear and high frequency batterychargers mentioned above may meet the benchmarks specified in the abovementioned regulations during certain operating conditions, such as fullload, the efficiencies of such chargers are below the specifiedefficiency benchmark at operating conditions other than full load. Thus,there is a need for a battery charger circuit that can meet theefficiency benchmarks set forth in the above mentioned regulations.

SUMMARY OF THE INVENTION

Briefly, the present invention relates to a hybrid battery charger whichincludes a linear charger circuit and a high frequency charger circuit.The hybrid battery charger takes advantage of the efficiencies of eachcharger circuit to improve the overall efficiency of the hybrid charger.The linear charger circuit is used to provide a high output current, forexample 75 amps even up to 300 amps, for vehicle starting applicationsand battery charging applications in which the battery charging currentis relatively high and exceeds a predetermined level representative ofrelatively high charging current, for example, over 7 amps. When thecharging current drops below the predetermined level, the linear chargercircuit is switched off and the high frequency charger circuit isswitched on to improve the overall efficiency of the hybrid charger. Inorder to further improve the efficiency of the hybrid battery charger,the frequency of the electronic switching circuitry in the highfrequency battery charging circuit may be varied to further reducelosses. In accordance with an important aspect of the invention, thehybrid battery charger circuit is fully operational even at relativelylow battery voltage conditions, for example, 1 volt DC.

DESCRIPTION OF THE DRAWING

These and other advantages of the present invention will be readilyunderstood with reference to the following specification and attacheddrawing wherein:

FIG. 1 is a block diagram of the hybrid battery charger in accordancewith the present invention.

FIG. 2A is an exemplary charging curve for a 12 ampere battery charger.

FIG. 2B is an exemplary charging curve for a 2.0 ampere battery charger.

FIGS. 3A-3G illustrate an exemplary schematic diagram of the hybridbattery charger in accordance with the present invention.

FIGS. 4A-4F illustrate an exemplary schematic diagram of amicrocontroller control circuit in accordance with the presentinvention.

FIG. 5 illustrates an exemplary schematic diagram of an alternateembodiment of a linear charger circuit in accordance with the presentinvention that utilizes a pair of SCRs instead of a triac.

FIG. 6 is an exemplary diagram illustrating the switching of a triac, oralternatively a pair of SCRs, used to control the magnitude of thevoltage from a 120 volt AC power supply that is connected to the linearcharger circuit.

FIGS. 7A-7L illustrate exemplary software flow charts in accordance withthe present invention.

FIG. 8A is a schematic diagram of the flyback transformer that formspart of the present invention.

FIG. 8B is a wiring diagram of the flyback transformer that forms partof the present invention.

DETAILED DESCRIPTION

The present invention relates to a hybrid battery charger which includesa linear charger circuit and a high frequency charger circuit. Thelinear charger circuit is used when it is most efficient; namely,providing a high output current, for example 75 amps, for vehiclestarting applications and applications in which the battery chargingcurrent, i.e., load current, is relatively high and exceeds apredetermined level representative of relatively high charging current,for example, over 7 amps DC. More particularly, in order to improve theoverall efficiency of the hybrid battery charger, the linear chargercircuit is switched off when the charging current drops below thepredetermined level, and the high frequency charger circuit is switchedon. Thus, the linear charger circuit and the high frequency chargercircuit are each used when it is most efficient to do so during acharging cycle.

Both battery charger circuits may be connected in parallel. As such, inapplications in which the required charging current exceeds the batterycurrent charging capacity of the linear charger circuit and the highfrequency charger circuit individually, both the linear charger circuitand the high frequency charger circuit are connected in parallel and canbe selectively turned on together to provide a charging currentequivalent to the sum of the maximum output currents of both chargercircuits.

In order to further improve the efficiency of the high frequency chargercircuit, a variable frequency drive circuit is used to control theon/off switching of the high frequency transformer. Typical drivecircuits are normally switched at around 100 KHz. The variable frequencydrive circuit allows the frequency of the switching to be controlledbetween 20 KHz and 100 KHz as a function of the output load conditions.In particular, during relatively low load conditions, the variablefrequency drive circuit reduces the switching frequency. At lowerfrequencies, the switching losses and thus power losses are greatlyreduced. Moreover, the variable frequency drive circuit can beconfigured to skip cycles at extremely low load conditions to furtherreduce losses. In the exemplary embodiment described and illustrated,the above mentioned improvements in efficiency enable the high frequencycharger circuit to attain efficiency over 80% for most load conditions.

The linear charger circuit, used for the engine start function andbattery charging, is capable of providing 10 Amps DC continuous batterycharging current. The linear charger circuit may be connected inparallel with the high frequency charger circuit that can provide 2 AmpsDC charging current. The net result is a hybrid charger that can provide12 Amps DC continuous charging current.

Since the linear charger circuit is only about 75% efficient, it canonly be used for a portion of the time in order for the hybrid batterycharger to attain an overall efficiency over 80%. In particular, inaccordance with one aspect of the invention, the linear charger circuitis only used when it is most efficient, for example, when one or more ofthe following conditions exist:

-   -   the charging current being drawn by the battery is above a        predetermined value, for example, approximately 7 Amps DC;    -   the battery voltage is below a certain value, for example, 13.2        Volts DC; and    -   the rate of change of voltage (dV/dt) is below a predetermined        value, for example, approximately 0.1V/hour.

By limiting the conditions when the linear charger circuit is used, asmentioned above, the linear charger circuit is only used when it is morethan 70% efficient. Using a charge time of, for example, 24 hours, asset forth by the Department of Energy in Section 430.23 of the Code ofFederal Regulations, the linear charger circuit is estimated to be usedfor 2 hours and would average an efficiency of approximately 75%.Assuming a worst case scenario in which the high frequency chargercircuit runs for an entire 24 hours, the average power used by thecombination of the linear charger circuit and the high frequency batterycharging circuit would meet or exceed 80% efficiency.

In order to meet the efficiency mentioned above, the hybrid batterycharger may include one or more of the following features to furtherreduce losses and improve efficiency. These features are optional andrelate to various aspects of the hybrid battery charger design.

One optional feature that may be incorporated into the hybrid batterycharger in order to further reduce losses of the hybrid battery circuitrelates to the linear transformer used in linear charger circuits. Inparticular, such linear transformers are known to dissipate power underno load conditions when connected to an external 120 volt AC powersupply. In order to further improve efficiency and reduce losses, thelinear charger circuit and thus the linear transformer may bedisconnected from the AC power supply when not in use.

Another optional feature to improve efficiency relates to currentlimiting the high frequency charger circuit. By current limiting thehigh frequency charger circuit, the efficiency is improved by keepingthe power supply from cycling between an ON condition and an OFFcondition when the battery is deeply discharged.

Another optional feature to further reduce power losses by the hybridbattery charger relates to the microcontroller used to control thehybrid battery charger. In particular, the hybrid battery chargerutilizes a microcontroller to control various aspects of the hybridbattery charger. For example, the microcontroller is used to makebattery voltage measurements; switch the linear charger circuit in andout; control the amount of starting current provided by the linearcharger circuit; and control the hybrid battery charger in accordancewith a charging algorithm. In order to further reduce power losses andincrease the efficiency of the hybrid battery charger, themicrocontroller may optionally be put into sleep mode when the chargegoes into maintenance mode. During a condition when the microcontrolleris in a sleep mode, the high frequency charger circuit is configured toregulate the high frequency charger circuit output voltage to themaintenance voltage, for example, 13.5V in response to an analog signalsince no signal will be available from the microcontroller during thiscondition. The microcontroller is set to wake under several conditionsincluding battery disconnection.

In accordance with another important feature of the invention, theconstruction of the high frequency transformer enables the charger tooperate normally at a battery voltage down to about one (1) Volt DC. Inparticular, a primary fly-back bias winding on the high frequencytransformer is wound on the bobbin first in multi-strands wire to coverthe full width of the bobbin in one layer, thus this bias winding worksas a shielding layer to improve EMC performance. In addition, the mainprimary winding is wound next and in opposite winding direction to thefly-back winding. This configuration improves the EMC performancefurther.

The high frequency transformer in accordance with the present inventionis provided with a second bias winding. One bias winding is connected inthe traditional fly-back configuration, i.e., connected with theopposite polarity as the primary winding, and the other bias winding isconnected with the same polarity in a forward direction. The reason thisis done is to allow the bias winding to supply voltage to run themicrocontroller and the PWM controller even if the load, i.e., batteryvoltage, is very low. The voltage of the fly-back bias winding isproportional to the battery voltage which is supplied by the outputfly-back winding. Thus, when battery voltage is very low (generallyknown as an over-discharged state, which may be 1 volt DC), themicrocontroller and PWM controller, which are powered by the fly-backwindings, will not operate correctly if those controllers are onlypowered by the fly-back bias windings, which are now very low due to thelow battery voltage. With help of the forward bias windings, which arein opposite polarity to the fly-back bias windings, those forward biaswindings then supply power to the microcontroller and the PWM controllerto keep them operating correctly.

Exemplary Block Diagram.

Turning to FIG. 1, a block diagram of the hybrid battery charger inaccordance with the present invention is illustrated. The hybrid batterycharger is generally identified with the reference numeral 20. Thehybrid battery charger 20 includes a high frequency charger circuit 22and a linear charger circuit 24. The high frequency charger circuit 22is selectively connected to the output terminals Vb⁺ and Vb⁻, generallyidentified with the reference numeral 25, by way of a switch SW1 28. Thelinear charger circuit 24 is selectively connected to an external sourceof 120 volts AC power supply 26 by way of a switch SW2 30. Both batterycharging circuits may be connected in parallel. In applications in whichthe charging current exceeds the charging capacity of the linear chargercircuit, both the linear charger circuit and the high frequency chargercircuit may be turned on simultaneously to provide a charging currentequivalent to the sum of the two charging circuits. More particularly,the switch SW1 28 allows the high frequency charger circuit 22 to beselectively connected to the output terminals 25 under the control of amicrocontroller 32, Similarly, the switch SW2 30 allows the linearcharger circuit 24 to be selectively connected to the 120 volt AC powersupply 26 under the control of the microcontroller 32.

The switches SW1 28 and SW2 30 allow the hybrid battery charger 20 tohave 4 modes of operation. In a first or starting mode of operation, thehybrid battery charger 20 operates to provide starting current, forexample, 75 amps DC. In this mode of operation, the switch SW1 28 isopen to disconnect the high frequency charger circuit 22 from the outputterminals 25. In this mode, the microcontroller 32 monitors the zerocrossings of the external source of 120 volt AC power. Depending on thelevel of starting current required, the microcontroller 32 will signalthe switch SW2 30 to close to deliver the required starting current. Forexample, with reference to FIG. 5, the nominal peak voltage of theexternal source of 120 volts AC. As will be discussed in more detailbelow, this voltage is applied to a step down transformer and rectifiedto provide a DC starting current and a DC charging current. Assuming astep down transformer with a turn ratio of 10:1, the voltage will bereduced by a factor of 10 and the current will be increased by a factorof 10. Assuming the external supply of 120 volts AC can supply 7.5 ampsof AC current, using the above example, the linear charger circuit 24 isable to provide 75 amps DC starting current.

In the starting mode of operation, the linear charger circuit 24 is usedto provide starting current. In this mode, the microcontroller 32 sensesthe zero crossings of the external source of 120 volts AC by way of azero crossing detector 34. Knowing the zero crossings and the frequencyof the external source of 120 volts AC, the microcontroller 32 is ableto trigger the switch SW2 30 at various voltage points along the 120volt AC sine wave by measuring the time from a zero crossing. In thisway, the linear charger circuit 24 is able to control the startingand/or charging currents. The shorter the time delay between a zerocrossing and the trigger signal to the switch SW2 30, the higher thestarting/charging current. More specifically with reference to FIG. 6,the various voltage points along the 120 volt AC sine wave correspond toinstantaneous voltage levels as a function of time. For example, thevoltage level V_(s) during the positive half cycle corresponds to thetime t_(s). Assuming that the time t_(s) provides the desired startingcurrent, the microcontroller 32 would trigger the switch SW2 30 to beclosed at the time t_(s) and the switch SW2 30 will reopen whenever thecurrent that flows through the switch SW2 30 has dropped to zero. Underthe control of the microcontroller 32, the desired starting current isapplied to the output terminals 25. In this mode of operation, theswitch SW1 28 is open, thus disconnecting the high frequency chargercircuit 22 from the battery during a starting current mode of operationto avoid back feeding the high frequency charger circuit 22. During thismode of operation, the microcontroller power Va is provided by thesecondary winding of the transformer T1 by way of the diodes D7 and D14.

In a second mode of operation, for example, a battery charging mode, thelinear charger circuit 24 is used to charge a battery during relativelyhigh load conditions, e.g., when the battery charging current is at orabove a predetermined level. For an exemplary 10 amp continuous linearcharger circuit, the predetermined level may be at or above, forexample, 7.0 amps DC. In this mode of operation, the switch SW2 30switches along the 120 volt AC sine wave at a relatively longer timedelay after detecting a zero crossing than the time delay of thestarting current mode to provide the required charging current under thecontrol of the microcontroller 32. In this mode of operation, the switchSW1 28 may be open.

In a third mode of operation, a high frequency charging mode, thebattery is charged by the high frequency charger circuit 22. In thismode, the switch SW2 30 is open, thus disconnecting the linear chargercircuit 24 from the circuit, and the switch SW1 28 is closed. Asmentioned above, the high frequency mode of operation is used when theload current, i.e., charging current is less than a predetermined level.A current sense resistor is placed in series with the output terminals25. The voltage across the current sense resistor 38 is read by themicrocontroller 32 as an indication of the charging current supplied tothe battery. Thus, when the charging current drops below a predeterminedlevel, as mentioned above, the hybrid charger 20 switches from thelinear charger circuit 24 to the high frequency charger circuit 22.

The hybrid charger 20 may be provided with an optional fourth mode ofoperation, parallel mode, in which the high frequency charger circuit 22is in parallel with the linear charger circuit 24. In this mode ofoperation, the linear charger circuit 24 and the high frequency chargercircuit 22 combine to provide a combined charging current that exceedsthe continuous rating of either the linear charger circuit 24 or thehigh frequency charger circuit 22 individually. For example, assumingthe high frequency charger circuit 22 has a continuous rating of 2.0amps DC and the linear charger circuit has a continuous rating of 10.0amps DC, the hybrid charger 20 can provide 12.0 amps DC continuouslywhen in a parallel mode of operation.

The hybrid battery charger 20 may optionally be provided with anoptional fifth mode of operation, a standby mode in which the switch SW128 is opened when the battery is disconnected from the output terminals25. In this mode of operation, the battery voltage is sensed by avoltage sense circuit 40. When the voltage sense circuit 40 senses aloss of battery voltage, the switch SW1 28 is opened under the controlof the microcontroller 32 to remove all power from the output terminals25.

In accordance with another important aspect of the invention, the hybridbattery charger 20 may incorporate a variable frequency drive circuit42. The variable frequency drive circuit 42 may be used to control theswitching frequency of the high frequency charger circuit 22. As will bediscussed in more detail below, the variable frequency drive circuit 42,for example, a “green mode” controller, such as a Fairchild SG6859A,controls the switching frequency as a function of the load current. Anexemplary variable frequency drive circuit 42 can be used to vary theswitching frequency of the high frequency charger circuit 22 from 20 KHzto 100 KHz, for example. At relatively low load conditions, the variablefrequency drive circuit 42 switches at the lower frequency in order toreduce switching losses. The variable frequency drive circuit 42 mayoptionally skip cycles when the green mode controller enters the greenmode, i.e., the voltage to the feedback pin of the PWM controller isabove 2.8 volts DC.

Schematic Diagram.

An exemplary schematic diagram of the hybrid battery charger 20including a control circuit is illustrated in FIGS. 3A-3G. An exemplaryschematic diagram of a portion of the control circuit that includes amicrocontroller 32 is illustrated in FIGS. 4A-4F. The schematic diagramof the hybrid battery charger 20 illustrated in FIGS. 3A-3G illustratesthe linear charger circuit 24 in which the switch SW2 30 is implementedas a triac. FIG. 5 illustrates an alternative embodiment of the linearcharger circuit 24 in which the switch SW2 30 is implemented as a pairof SCRs.

Linear Charger Circuit.

The linear charger circuit 24 (FIG. 1) is shown in FIGS. 3E-3G. Asmentioned above, the linear charger circuit provides starting current ina starting mode of operation and charging current in a charging mode ofoperation when the load current is above a predetermined value, forexample 7 amps DC, where the linear charger circuit 24 is mostefficient.

Turning to FIGS. 3A-3G, the linear charger circuit 24 includes a triacQ4 (FIG. 3F), which functions as the switch SW2 30, described above. Theinput to the triac Q4 is connected to the neutral terminal N of the 120volt AC power supply 26 (FIG. 3A). The output of the triac Q4 (FIG. 3F)is connected to one leg of a transformer T2 (FIG. 3G). The opposing legof the transformer T2 is connected to a terminal Ltx⁺ (FIG. 3F), whichis connected to the line terminal L of the 120 volt AC power supply 26(FIG. 3A) by way of a fuse F2 and a temperature sensing resistor RT2. Asurge suppressor V1 may be connected between the line and neutralterminals L and N, respectively, to suppress voltage spikes.

The transformer T2 (FIG. 3G), designed for working at 50/60 Hz (lowfrequency), is a current transformer for transforming the input currentapplied to the primary winding to provide either a starting current or acharging current as an inverse function of its turn ratio and thevoltage applied to the primary. As mention above, the triac Q4 (FIG. 3F)can switch at various voltage points along the 120 volt AC sine wave(FIG. 6) to provide either a desired starting current or a desiredcharging current.

The triac Q4 (FIG. 3F) is under the control of the microcontroller 32(FIGS. 4B, 4C, 4E, and 4F). More particularly, the microcontroller 32provides a series of phase control signals, identified as a “Phase Ctr.”signal. These control signals control the voltage point along the 120volt sine wave (FIG. 6) that the triac Q4 (FIG. 3F) switches, asdiscussed above. The phase control signals are applied to the triac Q4by way of a resistor R39 and an opto-coupler U5. The output of theopto-coupler U5 is applied to the gate of the triac Q4. A voltagedivider consisting of the resistors R37 and R28 is applied to the gateof the triac U5 to control the triac Q4 by way of the control signals. Acapacitor C21 is connected between the voltage divider and the input tothe triac Q4 to provide a steady signal to the gate of the triac Q4. Anoise filter consisting of a serially connected capacitor C18 andresistor R25 is connected across the input and output of the triac Q4 toprevent false switching due to voltage spikes in the 120 volt powersupply.

When the triac Q4 is conducting an AC voltage is applied to the primaryof the linear transformer T2 (FIG. 3G), as mentioned above. A pair ofrectifying diodes D22 and D23 is connected to the secondary winding ofthe transformer T2 to provide full wave rectification. During thepositive half cycle of the AC sine wave, the diode D22 conducts and thediode D23 is off. During this positive half cycle, a positive rectifiedDC output voltage is applied to the positive battery terminal Vb⁺.During the negative half cycle of the AC sine wave, the diode D23 is onand the diode D22 is off. Since the cathode of the diode D23 isconnected to the positive battery terminal Vb⁺, a positive rectified DCoutput voltage is still applied to the battery terminal Vb⁺ during thenegative half cycle of the sine wave to provide a continuous rectifiedDC voltage to the battery terminal Vb⁺.

The secondary winding of the transformer T2 is tapped and connected tosystem ground SGND. As shown, the secondary winding is shown with anexemplary center tap and connected to a connector CN1, which, in turn,is connected to the negative batter terminal Vb⁻.

An alternative linear charger circuit is shown in FIG. 5. In thisembodiment, two SCRs 04 and 08 are used in place of the triac. In thisembodiment, the output of the opto-coupler U5 is used to trigger thegates of the SCRs Q4 and Q8 separately. During positive half cycles, thediode D22 is off and the diode D25 and the SCR Q8 is conducting. Duringnegative half cycles, the diode D22 is on and the diode D25 is off andthe SCR 04 is conducting.

The SCRs 04 and Q8 are connected in parallel with a snubber circuit, asdiscussed above, consisting of the serially connected resistor R25 andcapacitor C18. A gate resistor R28 and capacitor C21 may be connectedbetween the opto-coupler U5 and the SCR 08.

As mentioned above, the linear charger circuit 24 is only used when theload current is more than a predetermined value, for example 7.0 ampsDC. While the linear charger circuit 24 is charging a battery, thecharging current is measured by the current sense circuit 38 (FIG. 1),as will be discussed below. The switch SW1 28 is closed and themicrocontroller 32 (FIGS. 4B, 4C, 4E, and 4F) reads the charging currentsupplied to the battery. When the charging current drops below apredetermined value, for example 7 amps DC, the microcontroller 32signals the triac Q4 (FIG. 3F) or alternatively, the SCRs Q4 and Q8(FIG. 5) to open by way of the Phase Ctr. Signal to disconnect thelinear charger circuit 24 from the 120 volt power supply 26 (FIG. 3A).As mentioned above, this is done so that the linear charger circuit 24(FIG. 1) is only used to charge batteries when the load current is abovea predetermined value, for example 7 amps DC, where it is mostefficient. Once the linear charger circuit 24 is disconnected from the120 volt AC power supply 26 (FIG. 3A), the high frequency chargercircuit 22 takes over and continues to charge the battery.

Current Sense Circuit.

The current sense circuit 38 (FIG. 1) is illustrated in FIGS. 3C and 3D.The current sense circuit 38 senses the battery charging current by wayof a current sense resistor R11A (FIG. 3C). One side of the currentsense resistor R11A is connected to the negative battery terminal Vb⁻(FIG. 3D). The other side of the current sense resistor R11A isconnected to ground OGND. The voltage across the current sense resistorR11A is applied to the current sense circuit 38 which includes anamplifier U4B, input resistors R24 and R30 and an output resistor R23.The ratio of the resistors R30/R24 determines the gain of the amplifierU4B. In this exemplary case, the gain is 20. The voltage across thecurrent sense resistor R11A is applied to the + and − terminals of theamplifier U4B. The resistor R30 and the capacitor C1 5, connected to thenon-inverting input of the operational amplifier U48 form a low passfilter for filtering input noise due to the high frequency switching ofthe voltage across the current sense resistor R11A. A pair of capacitorsC6 and C13 are connected between the output of the operational amplifierU4B and the input signal to stabilize the input and output voltages ofthe operational amplifier U4B.

The operational amplifier U4B amplifies the voltage across the currentsense resistor R11A and provides an output signal, identified asCURRENT. The CURRENT signal is connected to pin 8 of the connector CN1(FIG. 3G). The connector CN1, in turn, is connected to the connector CN2(FIG. 4A), which is connected to the microcontroller 32 (FIGS. 4B, 4C,4E, and 4F). As discussed above, the microcontroller 32 controls thecharging and starting current provided to a battery connected to itsoutput terminals Vb⁺ and Vb⁻ 25.

Power to the operational amplifier U4B is identified as a voltage Va,derived from the positive battery voltage Vb⁺ (FIG. 3D) by way of aresistor R26, diodes D7 and D14 and a resistor R7. A capacitor C14 isused to stabilize the voltage Va applied to a power supply input of theoperational amplifier U4B. The negative power supply input of theoperational amplifier U4B is connected to ground OGND.

Voltage Sense Circuit.

The voltage sense circuit 40 (FIG. 1) is illustrated in FIG. 3D. Thevoltage sense circuit 40 includes a transistor Q3. The emitter of thetransistor Q3 is connected to the positive battery terminal Vb⁺. Thecollector of the transistor Q3 is a signal identified as BATTERYVOLTAGE. This signal is applied to the microcontroller 32 (FIGS. 4B, 4C,4E, and 4F) by way of the connectors CN1 (FIG. 3G) and CN2 (FIG. 4A).The transistor Q3 (FIG. 3D) is under the control of the transistor Q2(FIG. 3C) and the diode D16. The diode D16 is connected to the negativepolarity side of one of the secondary windings of the flybacktransformer T1. As will be discussed in more detail below, the primarywindings of the flyback transformer T1 are switched on and off by aswitch, for example, a FET Q1. When the switch Q1 is closed, thenegative polarity side of the secondary winding of the flybacktransformer T1 will be positive, thus causing the diode D16 to conduct.The cathode of the diode D16 is connected to the base of the transistorQ2 by way current limiting resistor R8. A parallel combination of abiasing resistor R20 and a capacitor C20 is connected across the baseand emitter terminals of the transistor Q2 to bias the transistor Q2.This causes the transistor Q2 to be connected to the positive voltage onthe negative polarity side of the secondary winding and thus turn on.The collector of the transistor Q2 is connected to the base of thetransistor Q3 (FIG. 3D), which turns on the transistor Q3 (FIG. 3D). Theemitter of the transistor Q3 is connected to the positive batteryterminal Vb⁺. The collector of the transistor Q3 is the BATTERY VOLTAGEsignal which is applied to the microcontroller 32 (FIGS. 4B, 4C, 4E, and4F) by way of the connectors CN 1 (FIG. 3G) and CN2 (FIG. 4A).

Whenever the external 120 volt AC power supply is lost, for example,when the 120 volt AC is turned off by a user or the 120 volt AC isotherwise not available, the primary side circuit will lose power. Assuch, the PWM controller U1 will stop working and the FET 01 will turnoff. This causes the secondary windings W4 and W5 to lose power, which,in turn, will cause the transistors Q2 and Q3 to turn off. As aconsequence, the BATTERY VOLTAGE signal disappears, and themicrocontroller 32 knows the 120 volt AC power supply is lost andconsequently turns off the relay RLY1. This causes the bias power Va tolose power, which, in turn, causes all of the secondary control circuitincluding the microcontroller 32 to lose power and shut down. Inessence, the hybrid battery charger 20 stops working. Without the helpof a special circuit, which comprises a diode D16, a capacitor C20,resistors R8 and R20, transistors Q2 and Q3, and a resistor R1 2, themicrocontroller 32 will not know if the 120 volt AC power supply is lostsince the microcontroller 32 will get power from the batteryundercharging through resistor 26, diode D7 to the bias power Va eventhe 120 volt AC power supply has been lost. Thus, the underchargingbattery will eventually be discharged to dead when the 120 volt AC powersupply is unavailable.

Zero Crossing Circuit

The zero crossing circuit 34 (FIG. 1) is illustrated in FIGS. 3E and 3F.As mentioned above, the zero crossing circuit 34 determines the zerocrossings of the 120 volt AC power supply 26 in order to determine thecorrect point in time to fire the triac or the SCRs. The zero crossingcircuit 34 includes the diodes D11, D12, D17 and D18. The line L andneutral N rails are applied to the zero crossing circuit 34 by way ofcurrent limiting resistors R11 and R34. A capacitor C12 is providedbetween the line L and neutral N rails to filter the noise of the powersupply. During positive half cycles of the 120 volt AC power, the diodesD17 and D18 will conduct and during the negative half cycle, the diodesD11 and D12 will conduct. Resistors R11, R33 and R34 form a voltagedivider supplies AC voltage to the resistor R33. Whenever the voltage ofR33 is higher than the voltage of capacitor C17, a diode D19 willconduct causing a current to flow to the charging capacitor C17. Whilethe diode D19 is conducting, the transistor Q6 is off because its baseto emitter is reverse biased by the diode D19. Thus, during thiscondition, no current flows through the LED diode of the opto-couplerU3. As such, the phototransistor of U3 will be off and generate alogical “1” to the microcontroller 32. When the 120 volt AC power supplycrosses zero, the diode D19 is turned off, thus the voltage of capacitorC17 through resistor R33 is applied to the base of transistor Q6, thenturning on the transistor Q6 and the opto-coupler U3. When thetransistor Q6 is on during the zero crossing, a logical 0 is availableat the collector of the output transistor of the opto coupler U3indicating a zero crossing as indicated by the signal “Zero Crs.” Thesignal “Zero Crs” is applied to the microcontroller 32 by way of theconnectors CN1 (FIG. 3G) and CN2 (FIG. 4A).

As the AC voltage rises the diode D19 will be turned back on and thecapacitor C17 will continue to charge causing a logical “1” to beproduced at the output of the opto-coupler U3. At the next zerocrossing, the diode D19 again drops out and a logical “0” is provided asthe “Zero Crs” signal. The cycle repeats until the hybrid batterycharger 20 is disconnected from the 120 volt power supply.

High Frequency Charger Circuit.

The high frequency charger circuit 22 (FIG. 1) is illustrated in FIGS.3A, 3B, 3C, and 3D. The high frequency charger circuit 22 is connectedto the 120 volt AC power supply rails L and N by way of a fuse F1 (FIG.3A) and a NTC thermistor RT1 or other temperature sensor. An EMC filterconsisting of CX1 and an inductance L1 is connected between the line Land neutral rails N. A pair of serially coupled resistors R1 and R2 isconnected across the line L and neutral rails N in order to dischargethe capacitor CX1. A full wave rectifier consisting of the diodes D1,D2, D3, and D4 (FIGS. 3A and 3B) is also connected to the line L andneutral rails N. The full wave rectifier converts the 120 volt ACvoltage to a rectified DC voltage. A capacitor C2 (FIG. 3B) is connectedacross the output of the full wave rectifier to smooth out the DC outputvoltage of the full wave rectifier.

The high frequency charger circuit 22 includes a flyback transformer T1(FIG. 3C). In this case, the flyback transformer T1 includes a singleprimary winding, identified as W1, three (3) bias windings, identifiedas W2, W3, and W4 and a secondary winding, identified as W5. As will bediscussed in more detail below, the bias windings W2, W3, and W4 areseparated from the primary winding W1 by insulation, as shown in FIG.8B. The bias windings W2, W3, and W4 provide stable AC power to the PWMcontroller U1 and the microcontroller 32 during various operationalconditions including low battery voltage condition as discussed below.In particular, the negative polarity side of the primary winding W1 isconnected to the output of the bridge rectifier (FIGS. 3A and 3B). Acapacitor CY1 is used to improve EMC performance. The positive polarityside of the primary winding W1 is connected to primary side ground PGND(FIG. 3B) by way of a switch Q1 and a parallel pair of current sensingresistors R19A and R19B. The switch Q1, implemented as a FET, is used toconnect and disconnect the positive side of the primary winding W1 toground PGND under the control of variable frequency drive circuit 42(FIG. 1), discussed below. The variable frequency drive circuit 42causes the switch Q1 to switch between, for example, 20 KHz and 100 KHz,for example, as a function of the load current.

When the switch Q1 (FIG. 3B) is closed, the primary winding W1 isconnected to the DC output of the bridge rectifier. This causes primarycurrent to flow through the primary winding W1 which builds up themagnetic flux and energy in the transformer T1. During this condition, avoltage is induced in the secondary windings W4 and W5 (FIG. 3C) of thetransformer T1. During this time, a diode D8 (FIG. 3B), connectedbetween the negative polarity of the primary winding W1 and the outputof the bridge rectifier, is off. While the switch Q1 (FIG. 3B) isclosed, the diode D6 (FIG. 3C), connected to the positive polarity ofthe secondary winding W5, will also be off, thus preventing an outputcapacitor C4 from charging. The diode D13, diode D14 and D16 will be onbecause the windings W2 and W4 have the same polarity as winding W1.Thus, capacitor C7, C8 and C20 will be charged.

When the switch Q1 (FIG. 3B) is open, the diode D8 turns on to clamp thevoltage spike, caused by the leakage inductance of the transformer T1,to a safe level for the FET through the resistor R1 0 and the parallelresistors R5A and R5B and the capacitor C5. This causes positivevoltages across the bias winding W3 and the secondary winding W4, which,in turn, turns on the diode D5 (FIG. 3B) allowing the capacitor C8 to becharged to supply power to U1. This also turns on the diode D6 (FIG. 3C)allowing the output capacitor C4 to charge and supply load current tothe battery. The diode D7 is also turned on which provides power to therelay RLY1 by way of a resistor R31. With power to the relay RLY1, therelay contact is able to close allowing current to be delivered to thebattery when microcontroller 32 sends a signal to turn on the transistorQ5. A diode D15 may be connected across the relay RLY1 to act as asnubber during switching of the relay RLY1. The relay is under thecontrol of a relay control signal Relay Ctr by way of a resistor R32 anda switch Q5.

The output terminals Vb⁺ and Vb⁻ 25 are connected in parallel across theoutput capacitor C4, which supplies the load current to a battery cell(not shown) connected to the output terminals Vb⁺ and Vb⁻ 25 when theswitch Q1 is closed. When the switch Q1 is open, the secondary windingW5 through diode D6 charges the output capacitor C4 and supplies theload current to the battery as well. As discussed in more detail below,the on and off times of the switch Q1 are controlled by the variablefrequency drive circuit 42, discussed below.

Variable Frequency Drive Circuit.

The variable frequency drive circuit 42 (FIG. 1) is illustrated in FIGS.3B and 3C. The variable frequency drive circuit 42 may include a PWMcontroller U1, for example, a Fairchild Model SG6859A PWM controller forflyback converters. Other PWM controllers are also suitable. The PWMcontroller U1 includes a green mode function which decreases theswitching frequency under light load and no-load conditions. Thefeedback circuit 43 consists of a voltage feedback circuit and a currentfeedback circuit. The voltage feedback circuit is formed by the resistorR21, zener diode ZD1 and an opto-coupler U2. The voltage feedbackcircuit limits the Vin voltage to the predetermined value, for example,16.2V. The current feedback circuit is formed by the current sensingresistors R11A and R11B, the capacitors C3, C9, and C19, the resistorsR14, R22, R50, and R27, an operational amplifier U4A, a diode d10, andan opto-coupler U2. The opto-coupler U2 combines the voltage feedbacksignal and current feedback signal together, then sends it to the pin FBof the PWM controller U1. Once the feedback signal at pin FB of U1exceeds a threshold, the switching frequency decreases in order toconserve power during light load and no load conditions. For the abovementioned PWM controller, U1, the frequency can vary between 20 KHz and100 KHz. Decreasing the switching frequency dramatically reduces powerconsumption.

The current feedback circuit includes a difference amplifier U4A (FIG.3C) and an opto-coupler U2 (FIG. 3C) connected to a feedback pin FB(FIG. 3B) of the PWM controller U1. The difference amplifier U4A (FIG.3C) compares the voltage of the current sensing resistor R11A, whichrepresents the charging current with a reference signal. In this case,the reference signal is formed by the PWM Ctr signal, which comes fromthe microcontroller 32, through the RC filter circuit consisting of theresistor R27 and a capacitor C3. The output of the difference amplifierU4A is applied to the opto-coupler U2.

The PWM controller U1 (FIG. 3B) optionally includes a “green-modefunction,” for example, as provided by way of a Fairchild Model SG6859APWM controller or equivalent. The green mode function causes the PWM toautomatically reduce the frequency of the switching of the switch Q1 asa function of the load current. At relatively low load currentconditions, the frequency of the PWM controller U1 is reduced. Atno-load conditions, the frequency is reduced further by skipping anumber of switching cycles. Reducing the frequency and eliminatingcycles of the switching of the switch Q1 reduces the power losses.

As mentioned above, the high frequency charger circuit 22 has twocontrol modes, voltage control mode and current control mode. Before themicrocontroller 32 closes the relay RLY1 (FIG. 3C), the high frequencycharger circuit 22 works in the voltage control mode. In this mode, theresistor R21 is serially connected to zener diode ZD1. The serialconnection is connected between the output of the difference amplifierU4A and the positive battery voltage Vb+. Since the relay RLY1 is open,there is no current from the high frequency charger circuit 22 tobattery, the voltage of the amplifier U4A (FIG. 3C) is low, which causesdiode D10 to be off. Thus, only resistor R21 senses the voltage Vin andapplies the voltage feedback signal through ZD1 to opto-coupler U2.Whenever the Vin voltage is higher than the predetermined value, forexample 16.2V, the zener diode ZD1 will turn on, which in turn, turns onthe opto-coupler U2, which adjusts the voltage at FB pin of the PWMcontroller 42. In this way, the Vin voltage is maintained at thepredetermined value, so called voltage control mode. When the relay RLY1closes, a charging current flows to the positive battery terminal Vb⁺and returns from the negative battery terminal Vb⁻. The current sensingresistors R11A and R11 B sense the charging current and send the currentsignal to the non-inverting pin 3 of the amplifier U4A through the RCfilter which is consist of resistor R22 and capacitor C19. The amplifierU4A compares the charging current signal at pin 3 with a target chargingcurrent setting at the inverting input pin 2 of U4A which comes from thesignal marked as “PWM Ctr” from the microcontroller 32 through theconnector CN1, the RC filter which comprises the resistors R27 and R50and the capacitor C3. If the charging current signal at pin 3 of U4A ishigher than the target charging current, at pin 2 of U4A, which is setby the microcontroller 32, the output voltage at pin 1 of U4A becomeshigh, the diode D10 turns on, which, in turn, adjusts the voltage at FBpin of the PWM controller 42. Thus, the charging current is maintainedat the target charging current set by the microcontroller 32 when thecharger is in the current control mode. Since the voltage of a 12Vbattery is lower than the predetermined voltage Vin which is decided bythe voltage control mode, thus the voltage control mode is inactiveduring the current control mode. A capacitor C10 is also connected tothe feedback pin FB of the PWM controller U1 for voltage stabilization.

The PWM controller 42 has two different operation modes according to thevoltage level at its FB pin. They are “Green Mode” and “Normal Mode.”When the voltage at FB pin of the PWM controller 42 falls in a certainrange, for example, 2.2 volt 2.8 volt, the PWM controller enters “GreenMode.” Within the Green Mode, the operating frequency of the PWMcontroller declines linearly as the voltage at FB increases due to thecharging load declining. Thus the power losses at the switcher FET Q1,the transformer T1 and the output diode D6 decline as the PWM frequencydecreases, and it leads to higher efficiency at light load conditions.As the charging load increases, the voltage at FB pin of the PWMcontroller decreases. When the voltage at FB pin falls below a certainvoltage specified by the PWM controller 42, for example, 2.2 volts, thePWM controller 42 enters into “Normal Mode,” meaning the PWM controllerwill work at the maximum operation frequency predetermined by resistorR18.

A capacitor C19 (FIG. 3C) is connected to the non-inverting inputterminal of the difference amplifier U4A and ground OGND to filter thenoise from the charging current signal. Another capacitor C9 isconnected between the inverting terminal of the difference amplifier U4Aand the output and forms a negative feedback loop used to stabilize theoutput voltage of the difference amplifier U4A.

The power supply terminal Vcc (FIG. 3B) of the PWM controller U1 isconnected across the output of the bridge rectifier by way of a pair ofserially connected resistors R6 and R9 and a capacitor C8. The groundterminal GND on the PWM controller U1 is connected to the positive sideground PGND. A resistor R18 is connected to the R1 pin of the PWMcontroller U1 to create a constant current source and determine thenominal switching frequency. A current sense pin CS senses the voltageacross the resistor R19A and R19B through resistor R17, connected to thepositive polarity side of the flyback winding W3 through FET Q1 forover-current protection. The resistor R17 forms a voltage divider withthe resistor R15. The CS pin is also connected to ground PGND by way ofa capacitor C11. A gate drive pin GDR is connected to the gate of theswitch Q1 by way of a resistor R13 and parallel diode D9.

Low Voltage Operation.

The hybrid battery charger 20 allows normal operation down to very lowbattery voltages, for example down to 1.0 volts DC. In particular, thepower supply voltage for the microcontroller 32 (FIGS. 4B, 4C, 4E, and4F) is derived from a voltage, identified as Va (FIG. 3C). Duringconditions when the battery is deeply discharged, for example down to 1volt DC, the discharged battery essentially shorts out the secondarywinding W5 and dissipates the energy stored in the transformer. Duringconditions when the secondary winding is shorted out, the bias windingW3 will also effectively be shorted out because it has the same polarityas the secondary winding W5.

Normally, the voltage supply Va for the microcontroller 32 (FIGS. 4B,4C, 4E, and 4F) would normally be powered from the secondary winding W5by way of the diode D7 (FIG. 3C). When the secondary winding W5 isshorted by a deeply discharged battery, the voltage Va becomes too lowto maintain the supply voltage of 3.3 volt DC for the microcontroller 32to cause the microcontroller 32 (FIGS. 4B, 4C, 4E, and 4F) to likelyperform erratically. For the same reason, the bias winding W3 cannotmaintain a high enough voltage on capacitor C8 causing the PWMcontroller U1 to also not operate normally.

In order to solve this problem, a pair bias windings W2 and W4 with theopposite polarities to winding W3 and W5 are used to provide the properpower supply voltage to the microcontroller 32 (FIGS. 4B, 4C, 4E, and4F) and the PWM controller U1 (FIG. 3B) during a condition when thebattery is deeply discharged. In particular, the winding W3 (FIG. 3C) isconnected to a diode D5 (FIG. 3B) and a serially coupled resistor R4that will be coupled to the power supply pin VCC of the PWM controllerU1. Similarly, the winding W2 is coupled to a resistor R35 and a diodeD13 also coupled to the power supply pin VCC of the PWM controller U1.The power supply pin VCC of the PWM controller U1 is coupled between apair of serially coupled resistors R6 and R9 and ground by way of acapacitor C8. During conditions when the secondary winding W5 of thetransformer T1 is shorted out, the bias windings W2 and W4 still getpower from the primary winding W1, so that the PWM controller U1 andmicrocontroller 32 can keep working normally.

During the time the PWM drive pulse goes high, the FET Q1 turns onconnecting the non-dot terminal of the primary winding W1 to thepositive rail of the DC voltage at capacitor C2. This causes a currentflowing from the non-dot terminal to the dot terminal of the winding W1,which, in turn, induces currents flowing out of the non-dot terminals atwinding W2, W3, W4, and W5, respectively. Considering the polarity ofthe diode at each winding, only the diodes D13, D14 and D16, which areconnected to winding W2 and W4, respectively, will turn on to enablecharging of the capacitors C8, C7, and C20. The diodes D5 and D6, whichare connected to windings W3 and W5 respectively, will be off. Thewindings W2 and W4 are so called “Forward Windings,” while the windingsW3 and W5 are known as “Flyback Windings.” Since the diodes D5 and D6are off, the induced energy will be stored in the flyback windings W3and W5 during the period of the FET Q1 on. When the PWM controller U1turns off the drive pulse, the FET Q1 will turn off disconnecting Q1from the DC voltage at the capacitor C2. Thus, the forward winding SW230 and W4 will lose power also, and the diodes D13, D14 and D16 willturn off. Meanwhile the energy stored in the flyback windings W3 and W5during FET Q1 on, will force the diodes D5 and D6 to be forward biasedand turn on the two diodes, thus providing charging currents to thecapacitor C8 and C4 respectively. In summary, in one PWM cycle, duringthe PWM drive pulse high, the PWM controller U1 gets power from thewinding W2 through the resistor R35, diode D13 and the capacitor C8 andthe bias source Va to the control circuit. The microcontroller 32 getspower from the winding W4 through diode D14, resistor R7 and capacitorC7. During the PWM drive pulse low, the PWM controller U1 gets powerfrom winding W3 through the diode D5, the resistor R4 and the capacitorC8; while the bias source Va gets power from winding W5 through diodeD6, resistor R26, diode D7 and capacitor C7. Thus, there are two pathsto feed power to the PWM controller U1 and the bias power Varespectively. So even when the high frequency charger circuit 22 isshorted by deeply discharged battery, which means the PWM controller U1and the bias power Va will lose one power path which is through windingW3, diode D5, resistor R4 and capacitor C8 and through winding W5, diodeD6, resistor R26, diode D7 and capacitor C7 respectively, but they canstill get power from another path and keep working correctly.

The operational amplifier U4A provides current limiting. In other words,during conditions when the battery is deeply discharged, the operationalamplifier U4A exports a high signal to the opto-coupler U2 through thediode D10 and the resistor R14. If the charging current to the batteryis more than the value set by microcontroller 32, the voltage FB pin ofPWM controller U1 will reduce the PWM duty cycle, and the chargingcurrent is limited to the setting value.

As shown in FIG. 4A, the voltage Va is used to generate a 3.3 volt DCsupply voltage for the microcontroller 32 (FIGS. 4B, 4C, 4E, and 4F).The voltage of the regular U3 is controlled by the resistors R53 andR56. In this case, the regulator U3 generates 3.3 volts, which isapplied to the VDD input of the microcontroller 32 (FIGS. 4B, 4C, 4E,and 4F). The voltage Va, for example 15 volts DC, is applied to theemitter of a transistor Q6 (FIG. 4A). A bias resistor R26 is connectedbetween the base and emitter of the transistor Q6. The transistor Q6 isunder the control of a transistor Q11, which, in turn, is under thecontrol of the microcontroller 32 (FIGS. 4B, 4C, 4E, and 4F) by way of acurrent limiting resistor R34 (FIG. 4A) and a load resistor R31. Duringnormal operation, the LED display D3 and most of the LEDs (LED7-LED17)are off. To save energy consumption, the transistor Q11 and Q6 areturned off by microprocessor PC1, then the voltage regulator U3 is onlyconnected to Va through resistor R27. Since the value of resistor R27,for example, 1 KΩ, is relatively high, it limits the current to thevoltage regulator U3, in turn, saves energy consumption. When users wantto turn on the LED display D3 and other LEDs, the switches SW3 or SW4are depressed. The microprocessor PC1 needs more power from the 3.3Vvoltage regulator U3 to do so. If the 3.3V regulator U3 still only getspower from resistor R27, the power will be not enough to supply themicroprocessor PC1 and other LEDs. When the microprocessor PC1 detectsthat the switches SW3 or SW4 are depressed, it turns on the transistorQ11 (FIGS. 4B, 4C, 4E, and 4F). The transistor Q11 (FIG. 4A), in turn,turns on the transistor Q6, thus connecting the voltage Va to thecapacitor C7. Now the voltage Va is applied to an adjustable precisionshunt regulator U3 not only by way of the resistor R27 but also by of aresistor R32 which value is much smaller than resistor R27. Thus thevoltage regular U3 gets more power from Va to meet the need of morepower to the microprocessor PC1 and the LEDs.

Flyback Transformer Construction.

As mentioned above, the flyback transformer T1 includes a pair biaswindings W2 and W3, and bias winding W4. These windings are used toprovide power to the microcontroller 32 as well as the PWM controller U1during conditions when the battery voltage is relatively low, forexample, 1.0 volts DC. As discussed below, the windings are identifiedas set forth below.

Winding Name Type ½W1 NP1 Primary W2 NP3 Primary/Bias W3 NP4Primary/Bias W4 NS1 Secondary/Bias W5 NS2 Secondary ½W1 NP2 Primary

Exemplary construction drawings of the fly back transformer T1 areillustrated in FIGS. 8A and 8B. Referring first to FIG. 8A, the flybacktransformer T1 may include two (2) primary windings NP1 and NP2, twobias windings NP3 and NP4 and two (2) secondary windings NS1 and NS2.The bias winding NP4 is wound on first at one end of the bobbin inmulti-strands wire to cover the full winding width of the bobbin in onelayer, thus this bias winding works as a shielding layer to improve EMCperformance. In addition, the main primary winding is wound next and inopposite winding direction to this flyback bias winding NP4. It improvesthe EMC performance further. Since this bias winding NP4 is the onewhich most closest to the ferrite core of the transformer T1 and acts asa ECM shielding, it blocks the noise, created in the the primary windingW1, to reach the transformer core, in turn, reduces EMC noise. Further,the most noisy terminal 2 of the primary winding W1 is arranged to woundface to the most quiet terminal 5 which is connected to the ground PGND(FIG. 3B, 3C), this improves EMC performance further. All windings arewound with the polarities as shown in FIG. 8A. In this case, the primarywinding W1 is split into two portions, NP1 and NP2, which will beexplained in greater detail in the next section. In particular, onewinding NP3 is connected with the same polarity as the primary windingW1 and the other winding NP4 is connected in the opposite polarity.Similarly, secondary winding NS2 is connected with the same polarity asthe primary winding W1 and the other secondary bias winding NS2 isconnected in the opposite polarity. This configuration allows theprimary bias windings NP3 and NP4 and the secondary bias winding NS2 tosupply sufficient voltage to the PWM controller U1 and themicrocontroller respectively even if the battery voltage is low.

A winding diagram is shown in FIG. 8B illustrating an exemplaryconfiguration of the windings on a bobbin. As shown, the primary windingW1 is split into two portions, NP1 and NP2. A secondary winding NS1 issandwiched between the half primary winding NP1 and another half primarywinding NP2. In this way, the leakage inductance between the primarywinding W1 and the secondary winding NS1 is reduced largely, which, inturn, reduces the voltage spike to the FET Q1 when FET Q1 is turned off.An insulation tape may be disposed between the winding NP3 and thesecondary windings NS1 and NS2. The secondary windings NS1 and NS2 areconnected between the 9 and 10 and 9 and 7, respectively. An insulatedtape is disposed between the primary winding NP1 and the bias windingNP4.

Magnet wire is used for the primary windings NP1 and NP2 and the biaswindings NP3 and NP4. Triple insulated wire is used for the secondarywindings NS1 and NS2. The specifications for the transformer T1 are setforth below.

Material List

-   -   Ferrite Core    -   Bobbin    -   Magnet Wire    -   Triple Insulated wire    -   Insulation Tape

Winding Specification

-   -   NP4: 9 turns of 4×#32 magnet wire (4×=4 wires in parallel), one        layer, start 6 at far side→5.    -   NP1: 27 turns of 2×#29 magnet wire, two layers, start 2→3.    -   NS1: 9 turns of 2×#22 triple insulated wire, two layers, start        10→9.    -   NP2: 27 turns of 2×#29 wire, two layers, start 3→1    -   NS2: 4 turns of 1× triple insulated wire, one layer, spread out,        start 9→7.    -   NP3: 4 turns of 1×#32 magnet wire on the same layer as NP3,        start 5 (far side)→4.

Microcontroller Control.

The microcontroller 32 (FIGS. 4B, 4C, 4E, and 4F) may be a Model No.STM8S003K3T6 microcontroller, available from STMicroelectronics with 8Kbytes of flash memory, 1 kilobyte of RAM and 128 bytes of EEPROM. Themicrocontroller 32 controls a three digit display D3 (FIG. 4C) by way ofa plurality of transistors Q10, Q9 and Q7 and biasing resistors R33, R30and R28, respectively. The transistors Q10, Q9 and Q7 are under thecontrol of MUX signals MUX3, MUX2 and MUX1 which connect the anodes ofthe LEDs forming the display to a 3.3 volt DC. The cathodes of the LEDsforming the display are connected to the output ports are connected tothe output ports PD7, PD6, PD5, PD4, PD3, PD2 and PD0 on themicrocontroller 32 by way of the resistors R38, R39, R40, R41, R42, R43,R44 and R45, respectively. These output ports PD7, PD6, PD5, PD4, PD3,PD2 and PD0 control the operation of the LED display.

The microcontroller 32 also controls a number of LEDs as shown in FIG.4D. For example, the microcontroller 32 controls the LEDs; LED7, LED8,LED9, LED10, LED11, LED12, and LED13 by way of a transistor Q8, abiasing resistor R25 and an input resistor R29. A MUX4 signal from themicrocontroller 32 is applied to the input resistor R29 to connect a 3.3volt DC voltage to the anode of the LED7, LED8, LED9, LED10, LED11,LED12, and LED13. The cathodes of the LED7, LED8, LED9, LED10, LED11,LED12, and LED13 are connected to the output ports PD7, PD6, PD5, PD4,PD3, PD2 and PD0 on the microcontroller 32 by way of the resistors R38,R39, R40, R41, R42, R43 and R44, respectively. These ports PD7, PD6,PD5, PD4, PD3, PD2 and PD0 control the operation of the LED7, LED8,LED9, LED10, LED11, LED12, and LED13.

The microcontroller 32 also controls the LEDs; LED14 (FIG. 4F) LED18,and LED16. These LEDs are connected to output ports PC3, PC2, and PC1 onthe microcontroller by way of the transistors Q12, Q13 and Q14, loadresistors R48, R49, and R51 and the input resistors R50, R52, and R36.Similarly, the microcontroller 32 controls the LEDs; LED17 and LED15. Avoltage of 3.3 volts DC is connected to the anodes of the LEDs, LED17and LED15 by way of a pair of resistors R46 and R47, respectively. Thecathodes of the LEDs, LED17 and the LED15 are applied to the outputports PB4 and PB5, respectively.

The microcontroller 32 also monitors various switches, such as theswitches SW3 (FIG. 4A) and SW4. These switches SW3 and SW4 are connectedbetween ground and the output ports PB3 (FIG. 4E) and PB2, respectively.

A scaled amount of the battery voltage is applied to an input port PB0of the microcontroller 32. The battery voltage is scaled by a voltagedivider formed from the resistors R35 and R54. A diode D4 is connectedbetween the resistors R35 and R54 and 3.3 volts DC. A diode D5 isconnected in parallel with the resistor R54. This analog voltage at theport PB0 is then converted to a digital value by an analog to digitalconverter onboard the microcontroller 32. When the scaled value of thebattery voltage exceeds 3.3 volts, the diode D4 conducts so that thescaled voltage is clamped to 3.3 volts in order to protect the port PB0from damage by over-voltage. The diode D5 is to protect the port PB0from damage when a reversed polarity battery is connected to the chargercircuit, it also protect the microprocessor from damage by a staticelectric discharge.

The ground pin VSS is connected to the reset pin NRST by way of acapacitor C8 and connected to digital ground. The reset is held in ahigh logic state by way of a 3.3 volt DC voltage and a pull-up resistorR37. The VDD pin is the digital power supply pin. A 3.3 volt DC voltageis applied to the VDD pin. A capacitor C9 connected between the VDD pinand the digital ground stabilizes the input voltage to themicrocontroller 32. A capacitor C10 is connected between a regulatorcapacitor pin Vcap and digital ground. Programming of themicrocontroller 32 is by way of the SWIM port PD1.

Output control signals generated by the microcontroller 32 are set forthin the table below.

Signal Name Port Pin Number MUX3 PA3 7 MUX4 PF4 8 FAN Ctr PB6 10 RelayCtr PC6 23 Phase Ctr. PC5 22 PWM Ctr. PC4 21

Inputs to the microcontroller 32 are set forth in the table below.

Signal Name Port Pin Number Current PB1 15 Zero Crs PE5 17 BatteryVoltage PBO 16

Connections between the microcontroller 32 (FIGS. 4B, 4C, 4E, and 4F)and the electronic circuitry are by way of 10 pin connectors CN1 (FIG.3G) and CN2 (FIG. 4A).

The microcontroller 32 (FIGS. 4B, 4C, 4E, and 4F) has multiple low powermodes. For example, the microcontroller 32 may have wait, active haltand halt low power modes as set forth below. For a STMicroelectronics,Model STM8S003K3T6 microcontroller, these modes are defined in theirSTM8S003K3/STM8S003F3 application data sheet, DOCID018576, Rev.

3:

-   -   Wait mode: In this mode, the CPU is stopped, but peripherals are        kept running. The wakeup is performed by an internal or external        interrupt or reset.    -   Active halt mode with regulator on: In this mode, the CPU and        peripheral clocks are stopped. An internal wakeup is generated        at programmable intervals by the auto wake up unit (AWU). The        main voltage regulator is kept powered on, so current        consumption is higher than in active halt mode with regulator        off, but the wakeup time is faster. Wakeup is triggered by the        internal AWU interrupt, external interrupt or reset.    -   Active halt mode with regulator off: This mode is the same as        active halt with regulator on, except that the main voltage        regulator is powered off, so the wake up time is slower.    -   Halt mode: In this mode the microcontroller uses the least        power. The CPU and peripheral clocks are stopped, the main        voltage regulator is powered off. Wakeup is triggered by        external event or reset.

Another important feature is the ability to place the microcontroller 32(FIGS. 4B, 4C, 4E, and 4F) in a halt or sleep mode. In this mode, themicrocontroller 32 consumes the least amount of power. Themicrocontroller 32 can be woken up by applying a reset. This can be doneby applying a signal to the NRST pin by way of the connector J2 (FIG.4B). A switch press can be used to wake the microcontroller 32.Alternatively, a signal from an internal clock timer can be used.

The microcontroller 32 is programmed by a SWIM pin (FIG. 4C) on themicrocontroller 32. The SWIM pin on the microcontroller 32 is connectedto the SWIM pin 3 on the header J2.

Fan Control.

In order to provide cooling to the battery charger 20, a fan M2 (FIG.3D) may be provided. The fan M2 may be a 12 volt DC fan connected to Vinand ground OGND by way of a transistor Q7. A Fan Ctr signal is appliedto the input of the transistor Q7 by way of an input resistor R48.Anytime the Fan Ctr. signal is high, the fan M2 is turned on. A diodeD20 is connected across the fan M2 to protect the transistor Q7 from theback EMF generated when the motor M2 is switched off by providing acurrent path through the diode D20 and the motor M2.

Software Control.

Exemplary software control diagrams are illustrated in FIGS. 7A-7L. TheMain Loop is illustrated in FIG. 7A. Initialization of the hybridcharger 20 is illustrated by the block 100. On power-up of the hybridcharger 20, the system is initialized. Initialization includes settinginput/output ports, initializing the analog/digital converters,initiating the clock and the watchdog timer and initializing interruptsand system variables. After initialization, a burn in test is conductedin step 120. The burn in test entails simulating a battery charge for apredetermined period of time, for example 20-30 minutes, in order toscreen out problems not detected during a factory test.

After the burn in step, the system enters a loop 120 which includes thesteps 120-136. These steps 120-136 are continuously repeated while abattery is being charged.

Initially, the watchdog timer is reset in step 124. Next in step 126,the charge rate switch SW3 (FIG. 4) and the battery type switch SW4 areread by the microcontroller 32 to determine the selected battery typeand the selected charge rate. Next in step 124, the battery voltage isread from port PB0. After the battery voltage is read in step 124, thesystem checks the battery connection in step 130. The battery connectionis based on the battery voltage that is read in step 124. If the batteryvoltage is greater than a predetermined value, for example, 0.2 voltsDC, the system assumes a battery is connected to the output terminalsVb⁺ and Vb⁻ 25. If the voltage is less than 0.2 volts DC, the systemassumes no battery is connected and the system loops back to step 124and reads the battery voltage again. The system will loop between steps128 and 130 until a voltage of at least 0.2 volts Dc is read. Once thesystem determines that a battery is connected to the output terminalsVb⁺ and Vb⁻, the system proceeds to the charge handler in step 132 andcharges the battery. As indicated in the block 132, the charge handleris illustrated in FIGS. 7A-7J. After the charger handler routine isexecuted, the system determines in step 134 the status of the batterycharging by measuring the battery voltage and charging current andcomparing it with the values on the voltage and current nominal chargingcurves, for example, as illustrated in FIGS. 2 and 3 to determine thecurrent point on those curves of the battery and using that point todetermine the percentage complete of the battery charging. Once thebattery charge percentage is determined, it is displayed on the displayD3 (FIG. 4) in step 136 by a display handler, and the LED17 isilluminated to indicate a percentage complete of the battery charging isbeing displayed.

As mentioned above, the charge handler is illustrated in FIGS. 7A-7J.Referring initially to FIG. 7B, the charge handler is in an initialstate. In the initial state INIT state, the triac Q4 (FIG. 3F oralternatively SCRs Q4 and Q8 (FIG. 5) are turned off. The LEDs,LED7-LED18 (FIG. 4), are turned off. The fan M2 is also turned off. Thesystem then switches to a CHECK BATT state, as illustrated in FIG. 7B

Turning to FIG. 7B, once the system enters the CHECK BATT state, thebattery voltage is checked in step 140 to determine if it is greaterthan, for example, 0.2 volts DC. Each loop through the main loop, thebattery voltage is measured during a predetermined time period, forexample, 3 seconds. Thus, each loop through the main loop, a TotalCharge Timer or 3 second timer is initialized. If the battery voltage isless than, for example, 0.2 volts DC, a 3 second timer is reset and thesystem exits indicating that no battery is attached. If the batteryvoltage is greater than 0.2 volts DC, the system assumes a battery isconnected to the battery charger and illuminates the Connected LED 14.If the voltage is equal to or greater than, for example, 17.0 volts DC,as indicated by the block 144, the system assumes the battery is fullycharges and resets the 3 second timer in step 142 and then exits andilluminates the Charged LED 16. Alternatively, if the battery voltage isless than 17.0 volts DC, the system checks in step 146 to determine ifthe 3 second timer has expired which indicates that a new loop throughthe main loop is to be initiated, If the 3 second timer has not expired,the system proceeds to step 148 and turns on the Charging LED 18 (FIG.4), turns on the Fan M2 (FIG. 3) by way of the Fan Ctr. Signal,available at the PB6 port (FIG. 4) of the microcontroller 32. Thebattery voltage is saved as the variable Vorg. The system also clearsthe Total Charge Timer, i.e., 3 second timer and sets the appropriatecharge as set by the switch SW3 (FIG. 4A).

Three different charge rates are selectable by the switch SW3, namelySLOW, MEDIUM and FAST charge rates as well as START, which refers tostarting current. If a FAST charge rate is selected both the highfrequency charger circuit 22 (FIG. 1) and the linear charger circuit 24,are connected in parallel to the output terminals 25. Assuming thelinear charger circuit 24 can produce an exemplary 10 amps DC and thehigh frequency battery circuit can produce an exemplary 2 amps DC, afull 12 amps DC can be delivered to a battery in this mode, whichincidentally defines the fourth mode of operation discussed above.

During this mode, the relay contact RLY1 is closed connecting the highfrequency charger circuit 22 to the output terminals 25. In this mode,the microcontroller 32 provides a control signal “Relay Ctr.” whichcontrols a transistor Q5, which, in turn, controls the relay RLY1 so asto connect the linear charger circuit 24 to the output terminals 25during this mode.

During a FAST charge rate or fourth mode of operation, themicrocontroller 32 sends a “Phase Ctr.” signal to the triac Q4 (FIG. 3F)or alternatively the SCRs Q4 and Q8 (FIG. 5) at the appropriate time tosupply the proper charging current to the battery to be charged.

If a MEDIUM charge rate is selected by the switch SW3 (FIG. 4), only thelinear charger circuit 24 is used to charge a battery connected to theoutput terminals 25. During this mode, identified above as the secondmode, only the linear charger circuit 24 is connected to the outputterminals 25. The high frequency charger circuit 22 is disconnected fromthe output terminals 25 by way of a relay contact RLY1 under the controlof a relay RLY1, which, in turn is under the control of a transistor Q5.The transistor Q5, in turn is controlled by a signal “Relay Ctr.” fromthe microcontroller 32. During this mode, the linear charger circuit 24provides battery charging current to the battery to be charged, asmentioned above.

If a SLOW charge rate is selected, only the high frequency chargercircuit 22 is connected to the output terminals 25 during this mode ofoperation, identified above as a third mode of operation. During thismode of operation, the triac Q4 (FIG. 3F) or alternatively the SCRs Q4and Q8 (FIG. 5) are off disconnecting the linear charger circuit 24 fromthe output terminals 25, The relay contact RLY1 is closed in the mannerdiscussed above, connecting the high frequency charger circuit 22 to theoutput terminals.

If a START rate is selected by the switch SW3, the linear chargercircuit 24 is connected to the output terminals 25 in the mannermentioned above under the control of the microcontroller 32 (FIG. 1) toprovide starting current, for example 75 amps DC. for a predeterminedtime, for example, 5 seconds, under the control of the microcontroller32. During this mode the relay contact RLY1 may be open to disconnectthe high frequency charger circuit 22 from the output terminals 25during this mode.

Once the charge rate is selected, the system proceeds to the appropriatecharge algorithm, as mentioned above, and the appropriate charge rateLED is illuminated. The LED7 corresponds to a slow charge rate. The LED8corresponds to a medium charge rate while the LED9 corresponds to a fastcharge rate. The LED10 corresponds to starting current.

The system also checks the position of the switch SW4 with respect tothe battery type. Depending on the position of the switch SW4, the LEDcorresponding to the selected battery type will be illuminated. The LED12 corresponds to AGM batteries while the LED 13 applies to GELbatteries. The system then proceeds to the START CHARGE State, asillustrated in FIG. 7D.

In order to improve the overall efficiency of the hybrid charger, thelinear charger circuit 24 and the high frequency charger circuit 22, thesystem takes advantage of the conditions in which these charger circuitsare most efficient, for example, when one or more of the followingexemplary conditions exist:

-   -   the charging current being drawn by the battery is above an        exemplary predetermined value, for example, approximately 7 Amps        DC;    -   the battery voltage is below a certain value, for example, 13.2        Volts DC; and    -   the rate of change of voltage (dV/dt) is below a predetermined        value, for example, approximately 0.1 V/hour.

During conditions when the linear charger circuit 24 is in use, eitheralone or in conjunction with the high frequency charger circuit 22,i.e., FAST and MEDIUM charge rate conditions, the system monitorsvarious parameters, such as battery voltage, charging current and therate of change of voltage with respect to time, dV/dt. If any of theseparameters meet or exceed the values mentioned above, for example, thelinear charger circuit 24 is switched off, as mentioned above, andcharging is continued by the high frequency charger circuit 22 toimprove the efficiency of the hybrid charger. Turning to FIG. 7D, forthe selected charging rate, in step 150, the battery voltage is read andthe PWM duty cycle to the linear charger 24 is read to regulate thecharging current to the selected charging rate. Initially a first timer,for example, a 1 minute timer is initiated in step 152. During thisminute, a constant charging current is applied to the battery and thesystem checks for a “sulfation” condition. Battery sulfation is known tooccur when a battery will not accept a charge. As such, during the firstminute, the battery charging current is limited to a relatively lowvalue, for example, 1 amp DC. In step 154. After charging the battery ata constant current for a short time period, the system checks thebattery voltage to determine if it has increased. If not, the systemassumes the battery is sulfated in step 156 and switches to aDesulfation Mode in step 158 and exits. Alternatively, if the batteryvoltage increased during the low level charge, the system assumes thebattery is not sulfated, the system exits. After the first timer timesout, the system checks in step 160 whether the battery voltage isgreater than a predetermined voltage, for example, 10 volts DC. If thebattery voltage is less than the predetermined voltage. If not, thesystem checks a second timer, for example a 120 minute timer, in step162 to determine if the battery was charged at the limited chargingcurrent and the second timer timed out and the battery voltage was still<10 volts DC. If so, the system proceeds to an ABORT state in step 164.If the second timer has not timed out, the system exits.

If in step 160, the battery voltage is determined to be >10 volts DC,the system proceeds to step 166 (FIG. 7E). In step 166, the systemchecks the battery voltage to determine if the battery voltage is at itsnominal level. If so, the system proceeds to step 168. In step 168, thesystem saves the PWM duty cycle and sets a third timer, for example, a60 minute timer, and switches from a constant current mode to a constantvoltage mode. During a constant voltage mode, the system monitors therate of change of the charging current dI/dt and proceeds to a dI/dtstate.

Alternatively, if the system determines in step 166 that the batteryvoltage is not at its nominal value, V_(fish), the system checks in step170 whether the battery voltage is >a first predetermined value, forexample, 14.2 volts. If the battery voltage is greater than thepredetermined voltage, the battery voltage is checked at periodicintervals, for example, every 30 minutes in step 172. After everyinterval, the system checks whether the battery voltage has improved instep 174. If not, the system checks in step 176 whether the batteryvoltage has dropped below a second predetermined value, for example,14.7 volts DC. If not, in step 178, the system sets the variableV_(finish) to the predetermined value. The duty cycle is saved and thethird timer is set. The system also switches to the dI/dt state, asillustrated in FIG. 7F.

If it is determined in step 170 that the battery voltage is less thanthe first predetermined value, e.g., 14.2 volts, the system checks thebattery voltage at predetermined intervals, for example, every 120minutes, in step 180. After every interval, the progress is checked instep 182. If there is progress, the system exits and proceeds to thedI/dt state. If there is no progress, the system proceeds to step 184,which is the same as step 178.

The dI/dt state is illustrated in FIG. 7F. Initially, in step 186, thebattery voltage is regulated at V_(finish). The dI/dt is determined bychecking the pulse width of the charging current in order to regulatethe battery voltage at V_(finish). As such, the system checks in step188 whether the new duty cycle (DC) is <the old duty cycle. If not, thesystem checks in step 190 whether the new duty cycle is >greater thanthe old duty cycle+a predetermined value, for example 10. If not, thesystem checks in step 192 whether the third timer, i.e., 30 minutetimer, has timed out. If the system determines in step 192 that thethird timer did not time out, the system exits and proceeds to theMAINT_State. If the third timer has timed out, the system proceeds tostep 194. In step 194, the Charging LED1 8 (FIG. 4) is turned off andthe Charged LED 16 is turned on. In addition, the system proceeds to theMAINT_State.

The duty cycle in the constant voltage mode is also used to determine athermal runaway condition. In particular, if it is determined in step190 that the new duty cycle is >the old duty cycle+the predeterminednumber, i.e., 10, the system assumes a thermal runaway condition in step196. During this condition, the system turns off the Charging LED18(FIG. 4) and turns on the Charged LED16. The system then proceeds to theMAINT_State.

In step 188, if the system determines the new duty cycle is less thanthe old duty cycle, the system sets old duty cycle variable Old DC equalto the new duty cycle variable New DC in step 198. The system proceedsto step 200 and checks the charging current. Exemplary charging curvesare illustrated in FIGS. 2 and 3. As shown on these figures, the minimumcharging current is about 0.4 amps DC. The system compares the chargingcurrent with the minimum known charging current to determine if thecharging current is equal to or less than the minimum charging current.If the charging current is <than the known minimum charging current, thesystem assumes the battery is fully charged and proceeds to step 194. Ifthe charging current is not <the minimum current, the system exits andproceeds to the MAINT_State.

The MAINT_State is illustrated in FIG. 7G. During this state the batteryvoltage is regulated at V_(MAINT) in step 202 by applying a smallmaintenance charging current to the battery. In step 204, the systemchecks whether the maintenance charging current is greater than or equalto a predetermined value, for example, 1.0 amp DC. If not the systemexits and proceeds to a MAINT2_state. If the charging current is greaterthan the predetermined value, the system proceeds to step 206. In step206, the system initiates a fourth timer, for example, a 12 hour timer,and proceeds to the MAINT2_state.

The MAINT2_state is illustrated in FIG. 7H. In step 208, the systemcontinues to regulate the battery voltage at V_(MAINT) by applying amaintenance charging current. In step, 210, the system measures thecharging current to determine if it is greater than or equal apredetermined value, for example, 1 amp DC. If the maintenance chargingcurrent is determined to be greater than or equal to the predeterminedvalue, the system limits the maintenance charge current to thepredetermined value, i.e., 1 amp DC. The system continues charging thebattery at the limited maintenance charging current until the fourthtimer, i.e., 12 hour timer, times out as determined in step 214. If thefourth timer has not timed out, the system exits and proceeds to theDesulfation state. If the fourth timer has timed out, the systemproceeds to an Abort state in step 216.

The Desulfation state is illustrated in FIG. 71. Initially in step 218,the system checks the fourth timer to determine if more than 10 hours,for example, have elapsed. If more than 10 hours have elapsed, thesystem switches to the Abort state. If not more than 10 hours haveelapsed, the charging current is limited to a second predeterminedvalue, for example, 3 amps DC, in step 220. Next, the system checkswhether the battery voltage is less than a predetermined value, forexample, 13.8 volts DC. If the battery voltage is >than thepredetermined value, the system exits and proceeds to the Abort State.Alternatively, if the battery voltage is <than the predetermined value,the system proceeds to step 224. In step 224, the Charging LED18 (FIG.4) is turned off and the fourth timer, e.g., 12 hour timer is reset. Thesystem returns to the Start Charge state, illustrated in FIGS. 7D and7E.

The Abort state is illustrated in FIG. 7J. In this state, the triac Q4(FIG. 3) or alternatively, the SCRs Q4 and Q8 (FIG. 5) are turned off.In addition, the charging LED18 and the Charged LED16 are turned off.

A timer interrupt service routine for the Phase Ctr. PWM (FIG. 3) isillustrated in FIG. 7K. The interrupt service routine is used toperiodically determine the duty cycle of the charging current.Initially, all Phase Ctr. PWM interrupts are cleared in step 230.

The Phase Ctr. PWM interrupts are generated by the microcontroller 32(FIGS. 4B, 4C, 4E, and 4F) in order to sense the duty cycle of thecharging current. Assuming the linear charger 24 (FIG. 3F) is on, theduty cycle is obtained by the microcontroller 32 by measuring the pulseduration and the frequency of the charging current pulse train. The dutycycle is the ratio of the pulse duration to the pulse period. The pulseperiod is 1/pulse frequency. The duty cycle is determined in terms of acount or ticks of a timer.

In step 232, the variable Phase Ctr. PWM is set to be equal to the totalPhase Ctr. PWM count minus the duty cycle, as measured. In step 234 thevariable Phase Ctr. PWM count is incremented by 1. If the Phase Ctr. PWMcount is not less than the total count, as determined in step 234, thesystem assumes the battery is charged and proceeds to step 242 and turnsthe triac Q4 (FIG. 3F) or the SCRs Q4 and Q8 (FIG. 5) off.Alternatively, if the Phase Ctr. PWM count is less than the total PhaseCtr. PWM count, the system proceeds to step 238 to determine if thePhase Ctr. PWM count is greater than the low duty cycle LDC, the systemassumes the battery is not fully charged. As such, in step 240, thesystem turns on the triac Q4 (FIG. 3) or alternatively the SCRs Q4 andQ8 (FIG. 5).

FIG. 7L illustrates the zero crossing interrupt service routine forcontrolling the triac Q4 (FIG. 3) or alternatively the SCRs Q4 and Q8(FIG. 5), as discussed above. At every zero crossing the triac Q4 oralternatively the SCRs Q4 and Q8 are off, as indicated in step 242. ThePhase Ctr. counter is then reset in step 244.

Obviously, many modifications and variations of the present inventionare possible in light of the above teachings. Thus, it is to beunderstood that, within the scope of the appended claims, the inventionmay be practiced otherwise than as specifically described above.

What is claimed is:
 1. A hybrid battery charger comprising: a housing; aset of output terminals configured to electrically couple with abattery; a linear charger circuit positioned within said housing andconfigured to supply a first current to the battery via said set ofoutput terminals during a first mode of operation; a high frequencycharger circuit positioned within said housing and configured to supplya second current to the battery via said set of output terminals duringa second mode of operation, wherein the linear charger circuit iselectrically connected in parallel with the high frequency chargercircuit between a power source and the set of output terminals via oneor more switches; a variable frequency drive circuit positioned withinsaid housing and configured to control a frequency of the high frequencycharger circuit between 20 kHz and 100 kHz; and a microcontroller toselectively control said linear charger circuit and said high frequencycharger circuit via the one or more switches during a charging cycle ofthe battery, wherein the microcontroller is configured to switch thehybrid battery charger from the first mode of operation to the secondmode of operation during the charging cycle when a battery voltage ofthe battery measured across the set of output terminals is above avoltage threshold, and wherein the microcontroller is configured toswitch the hybrid battery charger from the second mode of operation tothe first mode of operation during the charging cycle when the batteryvoltage is not above the voltage threshold.
 2. The hybrid batterycharger of claim 1, wherein the variable frequency drive circuit isconfigured to control the frequency as a function of a load current atthe set of output terminals.
 3. The hybrid battery charger of claim 2,wherein the variable frequency drive circuit is configured to increasethe frequency as the load current at the set of output terminalsincreases.
 4. The hybrid battery charger of claim 1, wherein the highfrequency charger circuit and the linear charger circuit supply thefirst current and the second current to the battery simultaneously viasaid set of output terminals during a third mode of operation.
 5. Thehybrid battery charger of claim 1, wherein the first mode of operationis either an engine-starting mode of operation or a charging mode ofoperation, and the second mode of operation is a maintenance mode ofoperation.
 6. The hybrid battery charger of claim 1, wherein said highfrequency charger circuit includes a flyback transformer to providepower to said microcontroller during a low voltage condition.
 7. Thehybrid battery charger of claim 1, wherein the high frequency chargercircuit and the linear charger circuit supply the first current and thesecond current to the battery simultaneously via said set of outputterminals when a desired charging current exceeds a maximum limit of thefirst current.
 8. The hybrid battery charger of claim 1, wherein the oneor more switches comprises a first switch operatively coupled to saidmicrocontroller to selectively couple said linear charger circuit to thepower source.
 9. The hybrid battery charger of claim 8, wherein the oneor more switches further comprises a second switch operatively coupledto said microcontroller to selectively couple said high frequencycharger circuit to said set of output terminals.
 10. The hybrid batterycharger of claim 1, wherein the variable frequency drive circuit isconfigured to skip one or more switching cycles when a load current atthe set of output terminals is (i) zero or (ii) not detected.
 11. Thehybrid battery charger of claim 8, wherein the microcontroller openssaid first switch when said linear charger circuit is not charging thebattery, thereby prohibiting supply of power from the power source tothe linear charger circuit.
 12. The hybrid battery charger of claim 1,wherein the microcontroller is configured to enter a sleep mode during amaintenance mode of operation.
 13. The hybrid battery charger of claim12, wherein the high frequency charger circuit is configured to regulateits output voltage to a predetermined maintenance voltage in response toan analog signal.
 14. The hybrid battery charger of claim 12, whereinthe microcontroller is configured to exit the sleep mode when thebattery is disconnected from the set of output terminals.
 15. The hybridbattery charger of claim 1, wherein said high frequency charger circuitincludes a flyback transformer and said microcontroller includes a setof bias windings on a primary and a secondary of said transformerconfigured to provide power to a pulse-width modulation (PWM) controllerand the microcontroller during a low voltage condition.
 16. A hybridbattery charger for charging a battery, the hybrid battery chargercomprising: a housing having a display device positioned thereon,wherein the display device is configured to indicate a status of thebattery; a set of output terminals configured to electrically couplewith the battery; a linear charger circuit positioned within saidhousing to supply a first current to the battery via said set of outputterminals during a charging mode of operation; a high frequency chargercircuit positioned within said housing to supply a second current to thebattery via said set of output terminals during a maintenance mode ofoperation, wherein said high frequency charger circuit includes aflyback transformer, wherein the linear charger circuit is electricallyconnected in parallel with the high frequency charger circuit between apower source and the set of output terminals via one or more switches;and a microcontroller to selectively control said linear charger circuitand said high frequency charger circuit in accordance with said chargingmode of operation or said maintenance mode of operation during acharging cycle; wherein the microcontroller is configured to switch thehybrid battery charger from the charging mode of operation to themaintenance mode of operation via the one or more switches during thecharging cycle of the battery when a battery voltage of the batterymeasured across the set of output terminals is above a voltagethreshold, and wherein the microcontroller is configured to enter asleep mode during a maintenance mode of operation and to exit the sleepmode when the battery is disconnected from the set of output terminals.17. The hybrid battery charger of claim 16, further comprising avariable frequency drive circuit positioned within said housing andconfigured to control a frequency of the high frequency charger circuit.18. The hybrid battery charger of claim 16, wherein the microcontrolleris configured to control the high frequency charger circuit and thelinear charger circuit to supply the first current and the secondcurrent to the battery simultaneously via said set of output terminals.19. The hybrid battery charger of claim 16, wherein the display deviceis configured to indicate a charge percentage of the battery.
 20. Ahybrid battery charger comprising: a housing; a set of output terminalsconfigured to electrically couple with a battery; a linear chargercircuit positioned within said housing and configured to supply a firstcurrent to the battery via said set of output terminals during a firstmode of operation; a high frequency charger circuit positioned withinsaid housing and configured to supply a second current to the batteryvia said set of output terminals during a second mode of operation,wherein the linear charger circuit is electrically connected in parallelwith the high frequency charger circuit between a power source and theset of output terminals via one or more switches; a variable frequencydrive circuit positioned within said housing and configured to control afrequency of the high frequency charger circuit; and a microcontrollerto selectively control said linear charger circuit and said highfrequency charger circuit via the one or more switches during a chargingcycle of the battery, wherein the one or more switches comprises a firstswitch configured to electrically couple said linear charger circuit tothe power source and a second switch configured to electrically couplesaid high frequency charger circuit to said set of output terminal, andwherein the microcontroller is configured to switch the hybrid batterycharger from the first mode of operation to the second mode of operationduring the charging cycle when a charging current into the battery dropsbelow a current threshold.
 21. A hybrid battery charger comprising: ahousing; a set of output terminals configured to electrically couplewith a battery; a linear charger circuit positioned within said housingand configured to supply a first current to the battery via said set ofoutput terminals during a first mode of operation; a high frequencycharger circuit positioned within said housing and configured to supplya second current to the battery via said set of output terminals duringa second mode of operation, wherein the linear charger circuit iselectrically connected in parallel with the high frequency chargercircuit between a power source and the set of output terminals via oneor more switches; a variable frequency drive circuit positioned withinsaid housing and configured to control the high frequency chargercircuit at a frequency that is greater than 20 kHz, wherein the variablefrequency drive circuit is configured to skip one or more switchingcycles when a load current at the set of output terminals is (i) zero or(ii) not detected; and a microcontroller to selectively control saidlinear charger circuit and said high frequency charger circuit via theone or more switches during a charging cycle of the battery, wherein themicrocontroller is configured to switch the hybrid battery charger fromthe first mode of operation to the second mode of operation during thecharging cycle when a battery voltage of the battery measured across theset of output terminals is above a voltage threshold.